Modular audio amplifier front end

Inspired by Bob Cordell

Abstract

This is one of the amplifier front ends for the modular audio amplifier system. This front end is heavily inspired by Bob Cordell's MOSFET amplifier with error correction and his excellent book about designing audio power amplifiers. I learned a lot from his book and tried to apply that in this design, plus some further great ideas from other designers and rather unusual design decisions on my own.

The design is exceptionally complex for a DIY project and development of the module turned out to be significantly more work than anticipated until I was rewarded with both a well working amplifier and a lot of experience gained.

Features

  • Extremely high power supply rejection
  • Power supply invariant operation with wide operating range
  • Input over-voltage protection
  • Negative feedback loop stays intact during heavy clipping
  • Adaptive clipping by sensing the output stage power supply
  • Many options for advanced compensation techniques like MIC and TPC

Schematic

The schematic looks more complicated than it actually is. The first page shows all circuits that are not in the signal path. The second page is basically the front end of Bob Cordell's amplifier with some nice extras added.

Power Supplies

The front end requires two floating 15V supplies in order to work properly. Without extra supplies for the front end, output signal voltage headroom is pretty limited. The floating supplies are added on top of the power amplifier supply rails and are then filtered by capacitance multipliers. Bob's design already features high and symmetric PSRR so extra filters are not necessary, but additional filters do no harm and have further benefits than just improve PSRR significantly. Power supply decoupling after the active filters can not pollute the ground with noise. Also, subsequent stages, which may not have such excellent PSRR as the front end, being fed from the filtered power supplies cannot degrade PSRR of the amplifier.

The downside of the active filters is that the front end power supplies ramp up slowly, which may lead to some oscillation during the ramp-up phase.

In case active filters are not desired, it is easy to form a passive filter instead by different component assembly. An example for the positive supply filter would be to leave away Q101, Q103, D103 and R105, install 0Ω resistors in place of R103 and D105, and install a low value resistor instead of D107 and D108 (maybe 10Ω).

The schematic shown here is suitable to build amplifiers of any power rating. It will run fine with output stage power supply rails as low as +/-12V and as high as +/-85V. For a very high power design, I would change some resistor values to reduce dissipation. For supplies over roughly 60V, all 20kΩ resistors should be roughly 27kΩ or 33kΩ. Bias is almost supply voltage invariant because the output of the CCS used in this design are nearly independent of supply voltage. Radiators are optional and can be left away for low supply voltage operation.

Power Supply References

The rail voltage reference circuits derive a filtered reference mixed from both supply rails, which is used as reference for the feedback Baker clamp circuitry. This adaptive reference in combination with the feedback baker clamps ensure that the front end and subsequent stages cannot be driven into saturation. The reference voltage generated ensures that the reference is lower than the supply voltage of the output stage and lower than the cascode reference voltage. Even in case of severe overdrive, the front end remains linear with the feedback loop intact.

By design, the adaptive voltage references cannot have good low frequency PSRR.

Current Sources

Four constant current sources bias the front end: One for each of the long tailed pairs and two are just for the cascode reference of the first LTP. The tail CCS design is based on my previous investigation of constant current sources for audio applications. The two CCS for the cascode reference are simple cascoded CCS.

I added a CCS to the second LTP because this makes the second LTP standing current independent from the first LTP current. The CCS used for the first LTP is very stable, but having the ability to set both currents independent from each other makes experiments much easier. Also, the output signal is now current limited, which might be seen as advantage or disadvantage dependent on point of view. I see current limiting as an advantage because it helps to prevent destruction in case of some potentially destructive events. The cost of the extra CCS is more voltage headroom required. However, the front end requires boosted power supply rails anyway so spending the extra voltage here is no loss.

Audio Input Filter

The input filter offers many options: C3 is a huge film capacitor, which is my preference. As an alternative, a bipolar electrolytic capacitor (C4) can be installed with a small film capacitor (C5) in parallel optionally. Resistor R3 ensures that no DC builds up. R2 and C2 may be used to properly terminate the audio input cable. They should only be installed in case a long cable connects to the amplifier input directly and be omitted in case any pre-amp or audio input transformer is used. R4, R5 and C6, C7 form a second order low pass filter with a corner frequency of 300kHz. Most amplifiers use a first order filter here, but I find that a second order filter has better performance.

Resistors R4+R5+R6 sets the input impedance. I chose 30kΩ because I may want to feed the amplifier from an audio input transformer and lighter loading of the transformer is preferable. Resistors R4 and R5 could be chosen lower in value and C6 and C7 higher in turn. Again, the reason for the values chosen here is to maintain compatibility to the audio input transformer I might want to connect.

R1 and C1 are the usual measures to break ground loops and may be replaced by a direct connection instead.

Audio Input Protection

The amplifier input protection I developed earlier, is implemented here. This is supposed to be a robust amplifier building block. In case no protection is required or desired, the components simply can be left away.

Complimentary Feedback Pairs in the input long tailed pair

The input LTP allows to use CFPs instead of single transistors. This improves linearity of the LTP. I like to build it this way since more inherent linearity prior to the application of negative feedback is a good idea in my opinion. Also, the LTP is degenerated a bit more than in most designs to further increase linearity. According to Douglas Self, CFPs in the LTP may improve distortion up to eightfold when regarding the LTP on its own, but in a complete amplifier, the improvement is mostly in the high frequencies. A tail current of 8mA ensures high transconductance and slew rate are maintained. Since the LTP transistors are cascoded, a broad range of transistors can be used. Transistors Early effect is also reduced by cascoding. In case the CFP should not perform as desired, Q23 and Q24 can be removed and R13 and R14 be bridged in order to fall back to a normal LTP design.

One minor downside of the CFPs here is that this costs a tiny bit of phase margin. Negative feedback becomes less effective at higher frequency, while the inherent linearity is still there.

Hugh from AKSA confirmed that CFPs inside LTPs work well an he has successfully done that before in his amplifier designs. The CFPs linearize the gm of the active devices so that the horizontal S curve is flattened. This is especially effective in for high amplitude signals. So after I took the decision to use CFPs, I learned that this was a good idea and why.

The CFP arrangement also reduces memory distortion. I recommend to study the work about memory distortion that Pierre Lacombe has presented.

Differential driven cascode

The first LTP features two independent cascode reference voltages that are derived from the LTP emitter potential of each input transistor. The benefit of having separate cascode reference voltages for each leg of the LTP driven from the emitter potential of the LTPs transistors is that VCE of each transistor (Q1 to Q4) is perfectly constant regardless of the signal level. This lowers any distortion associated with varying VCE across the transistors.

Previously, I saw proposals to drive the common cascode reference with a scaled down replica of the output signal, but was not excited by the idea to feed the input from the output. Also not encouraging was the observation that simulation did not show any significant advantage of a simple driven cascode, but this turned out to be a fallacy. According to other designers observations, it seems that the VCE parameter dependency is not modelled well in many transistor models and therefore the advantage shows in reality, but not in simulation.

The driven cascodes were added as refinement in the third revision of the schematic. Previously the cascode reference was generated by feeding a constant current from the positive power supply trough a string of LEDs and the current being dumped into the tail CCS. The part count of the driven cascodes is nearly the same considering that the cascode reference CCS is of the same design like the tail CCS.

Floating cascode

I added transistor Q19 as floating cascode into the second LTP mainly for distributing power dissipation in case of high voltage application. Q19 could be removed and C-E shorted in case this is not desired. However, cascoding the cascode makes the load more symmetric, which I believe is a good decision.

Feedback Baker Clamp

The feedback baker clamps are explained in Bob's book and I can't explain this topic nearly as well so I refer to the book at this point. The basic idea is to avoid saturation of the amplifier by diverting excessive signal swing beyond a certain threshold back to the inverting input of the amplifier where it combines with the rest of the (clipped) signal. This way the feedback loop stays intact even under heavy clipping conditions. Saturation and recovery issues of transistors are therefore avoided.

Components and substitutes

At the time of writing, some of the transistors in the schematic are obsolete. This does not really matter since the design does not seem to depend on specific transistor types too much.

All BC547C / BC557C transistors stand for just any high hFE and high fT complimentary small signal transistors that may be available. Transistors being complimentary is nice, but not that important in this application.

All BC550C / BC560C stand for low noise transistors can be substituted by any complimentary low noise transistors that may be available. Transistors being complimentary is nice, but not that important in this application.

The SC3503/SA1381 transistors are unique, but the design also simulates well using the TTA004B and TTC004B transistors. Using those transistors instead results in a tiny loss of phase margin. Also, the SOA of the TTA/TTC transistors is much lower and this limits the possible supply rail voltage. I use both types in the schematic to indicate which ones are preferred to have low Cob like the SC3503/SA1381 feature.

The MJE340/MJE350 transistors were chosen due to their excellent SOA.

In general, for low power designs, lower power transistors could be used that are easier to find having good specifications. The components in the schematic and their operating conditions shown here are suitable for amplifiers up to several hundreds of Watts output power.

Compensation

Miller Input Compensation

Miller input compensation is the original compensation scheme Bob Cordell used for this front end in the eighties. The input stage is included in the compensation loop. The big advantage of this compensation technique is that very high slew rates can be obtained because the current into the Miller compensator is no longer limited by the input LTP CCS. Also, the LTP distortion is reduced by inclusion into this loop. The downside is that this loop is much more difficult to stabilize. I wanted to keep much of the Cordell spirit alive and so I kept this compensation scheme. I found getting the amplifier stable very challenging and it took a good deal of simulation and experimentation to obtain a somewhat stable amplifier.

In the meantime, Bob would favour some different ways of compensation like TMC, maybe combined with MIC. For my amplifier, I don't see a big advantage of including the first LTP in the loop because I added CFPs to linearize the LTP, which are effective without using risky compensation techniques.

The difficulty is that six components are involved in this compensation technique. This is the most complex compensation technique known to me.
Simulation is only partially helpful because this results in a vast variety of setups that appear stable in simulation, but oscillate terribly on the bench. In case the compensation is off, the amp shows sustained several MHz and high amplitude oscillation in idle condition and any other condition. I smoked several components until I finally succeeded.
When iterating compensation on the bench, the question is where to start at all. Bob's book (first edition) does not explain how to calculate or iterate Miller input compensation. I found some guidelines Bob has shared on DIY Audio that I copied here for reference with component names adjusted to my schematic so that it makes sense in this context:

Let's assume I've chosen a closed loop gain of 20 and a gain crossover frequency of 2MHz. This means that I want the forward gain of the combined input stage and VAS to be 20 at 1MHz, assuming that the output stage has unity gain (if the output stage had a gain of 2, the target would be 10 instead). Having chosen the feedback shunt resistor (R64 + R65) to be 215 ohms, I now know that the reactance of C81 must be about 19 times 215 Ohms at 1 MHz, or about 4k. Thus C81 becomes 20pF.
I then want to put in a stabilizing zero at a frequency above the gain crossover frequency. This is done with R81 at 680 ohms. This gives me a zero at about 11 MHz.
Next I analyze and stabilize the inner loop that is closed by C81 and R81. This can be a high-bandwidth loop with a gain crossover beyond 10 MHz, since only small-signal fast transistors are in the loop. This fast loop is stabilized by C80 and R80. This is very light lag compensation.
Note that there is not a lot of interaction at the base of Q2 because the 215 ohm resistor is small compared to the impedances of C81 and R81.
The advantage of the scheme is that it rolls off the high frequency forward path gain by applying feedback around the input stage, increasing its dynamic range, reducing its distortion, and not making it work harder at high frequencies (the way Miller compensation does). It only has to work harder at very high frequencies where C80 comes in, but this is pretty far out.
The ULGFs of the major and minor loops are so far apart, perhaps a decade, that the major loop feedback contribution to the minor loop gain of the MIC is pretty much inconsequential in the frequency range of the minor loop ULGF.
Compensating the MIC minor loop can be tricky because there are so many possible combinations of the resistance and capacitance values. However, a simple rule-of thumb approach can be used to converge on a circuit fairly quickly. There are three R-C pieces to the design. The first is C81, the MIC dominant pole compensation capacitor against feedback resistor R60 to R63 and feedback shunt R64 + R65 (ignore R81 for now and let it be zero, as it often can be). The global ULGF is set by making C81s reactance equal to R60+R61+R62+R63 at the desired global ULGF.
The second R-C piece is the IPS shunt lag-lead load consisting of C80 and R80. This is the primary compensation network for the MIC loop. For now, you can assume R80=0Ω.
The third piece is the VAS R-C load consisting of C82 and R82. In some designs you can get away without it, but I strongly recommend it. It establishes a predictable and dampening load for the VAS at high frequencies. It also controls what proportion of the VAS signal current flows back through C81. Indeed, for most designs you can arbitrarily set C82 to 50pF. If you make it too big, you will impair the achievable slew rate of the VAS. Now, select R82 to set the time constant of C82*R82 to be the same as the time constant of C81 and the resistance it has in series with it. The division of VAS signal current between the two paths (through C81 and C82) will now be largely flat with frequency.
Now just set the value of C80 (I just do it by simulation) to achieve the desired minor-loop ULGF. I usually choose about 10MHz. Optionally add some resistance at R80 to optimize phase and gain margin of the minor loop. Resistance can be added at R81 to add some phase lead to optimize the global gain and phase margin, if desired. If resistance is added at R80, one may wish to appropriately revise the value of R82. Note that even with R81 = 0Ω, the non-inverting amplifier structure of the MIC loop prevents its gain from going below 0dB, so a zero is created there no matter what.
With R81=1k, the forward open-loop gain of the compensated amplifier cannot go below +6dB (i.e., X2) in theory. If the closed loop gain of the amplifier is 20, and ULGF is 1MHz, the zero will occur where the closed loop gain will have fallen to 6dB at 6dB/octave. This will be one decade above 1MHz, i.e., 10MHz.
If there is peaking in one of the local loops, it can cause peaking in the overall open loop gain, and this will inevitably result in peaking in the closed loop gain at frequencies above the ULGF.

I found that all six components are required and need to be optimized. In case any component is not optimized, the amplifier will oscillate full bore.

There are further challenges with MIC that need to be solved. When using BJTs in the input LTP instead of JFETs, the gm is higher, which requires either more input LTP emitter degeneration to lower gm or higher value for compensation capacitor C80. Increasing C80 leads to excessive charge storage in the compensation capacitor during clipping when the LTP is over-driven unless the amplifier is prevented from clipping (by using feedback Baker clamps for example). Increasing the second LTP emitter degeneration was observed by Glen Kleinschmidt as possible way to help deal with this effect.

Most of my frustration with MIC, apart from the sheer complexity of this compensation scheme, stems from unrealistic expectations regarding possible UGLF given the size of my power output stage. The size of the output stage ultimately limits the attainable UGLF and it took me quite a while to realize that four pairs of huge BJTs in parallel simply don't go well with 1MHz UGLF. Since MIC has the typical 6dB per octave roll-off, either the UGLF needs to be lowered accordingly or the output stage being shrunk accordingly.

Miller Compensation

For simple Miller compensation C84 with C85 in series can be used. This was an addition with revision 2 of the design as fallback measure in case I would not succeed getting the amplifier stable with Miller Input Compensation.

Two Pole Compensation

Two pole compensation makes the gain roll off at 12dB per octave instead of 6dB in normal Miller compensation and therefore shows impressive feedback up to high audio frequencies. It is very important to return the 12dB per octave slope back to 6dB per ocatave prior to reaching UGLF in order to maintain stability. The midband phase droop is of less concern, but the phase margin at UGLF is important for stability. C85 is the Miller capacitor in the two pole compensation and C84 should have (at least) two times to ten times the capacitance of C85. The ratio of C84/C85 determines the amount of additional open loop gain below unity gain crossover frequency. The lower the ratio, the higher the gain, but the less stable is the amp. With the capacitors fixed, R84 sets the frequency of the zero and therefore the point where the gain slope transitions from 12dB per octave to 6dB per octave.

The shunt network R80 / C80 is obsolete with TPC since capacitor C86 shunts the signal already.

Douglas Self's Book "Ampliier Design Handbook" is most helpful getting started with, understanding and optimizing TPC. This is what I would recommend to study first. And I found the AES convention paper by Harry Dymond and Phil Mellor very good as in depth study of TPC. The paper from Michael Kiwanuka is a bit difficult to digest.

TPC mid-band gain peak

With an additional capacitor added in parallel to the two serial connected compensation capacitors, the peak in loop gain typical for two pole compensation can be flattened. Bob Cordell calls this bridged T compensation and suggests that this capacitor should have roughly a tenth of the combined value C84 and C85 have. In simulation I found much smaller values more appropriate. The difficulty with such small capacitance values is that any minor disturbance from the PCB design may have big influence. In revision two of the PCB, I added a footprint for such a capacitor, that I revoved later in revision three of the PCB.

Douglas self proposes an alternate way to get rid of the mid-band peak in the loop gain by adding a resistor parallel to C85. The value of this resistor may be 1MΩ to 2MΩ roughly. I find this more elegant and this is my preferred solution to flatten the mid-band gain peak. R85 serves this purpose in my schematic.

Harry Dymond proposes to add a capacitor in series with R84 (P4) to both improve the mid-band phase droop and the open loop gain peak at once. Read the AES paper for details. I decided to not try this solution, mainly due to the PCB being overcrowded in this region.

TPC closed loop gain peak

Douglas Self describes in his book how the value of the TPC resistor also affects a closed-loop gain peak, which is usually located at 100kHz to maybe 300kHz. The lower the resistor, the higher the gain peak and the higher the gain peak frequency. I observed this effect and it even manifested in tendency to oscillation at the gain peak frequency.

Assembly in CAD

The front end PCB is unusually large. Many amplifier designs fit all stages onto smaller PCBs. With the size of the schematic in mind it is no surprise that the PCB is that large. Placement is pretty dense in order to minimize circuit loop area. Circuit blocks are well separated from each other to minimize interference. From the placement area overview it becomes clear that the actual amplifier circuitry occupies less than half of the PCB area. Of course, all mounting holes are located on the 20mm grid of the modular audio amplifier chassis.

Simulation

PSRR

Setup for simulation is with the non-inverting input grounded, a voltage controlled voltage source as substitute for the ouput stage and the AC voltage source for injection of the disturbance placed in series with either DC voltage source.

The compensation scheme plays an important role in PSRR.

PSRR with MIC

Miller Input Compensation

Following plots were made with the front end being compensated using Miller input copmensation, which helps to yield very high PSRR.

Simulation of power supply rejection shows inherently good and somewhat symmetric rejection of both power supply rails for the Cordell front end without any additions.

The addition of R-C filters of 10Ω and 82µF for the front end supplies help to avoid loss of power supply rejection at higher frequency. This shows how much effect a few cheap components can have. At this point, return of investment regarding further improvement becomes somewhat questionable.

However, since I need boosted power supply rails anyway, I like to turn the additional voltage headroom into something useful and this is why I added active filters. With the active filters, power supply rejection is further improved considerably. Grunge from the power supplies does not contribute to most genres of music so I prefer to minimize this as far as possible. Most recordings unfortunately feature enough annoying background noise already.

Low frequency PSRR is worsened by the adaptive rail voltage references, but in combination with the active power supply filters, this effect is compensated mostly. I still haven't understood why the adaptive voltage references worsen PSRR at all because the reference voltage is only relevant in case of clipping the signal. It seem un-intuitive to me that there is an effect during normal operation. In any case, PSRR is still acceptable.

PSRR with TPC

Two Pole Compensation

With TPC, the PSRR is by far not as good as with MIC. An important consideration here is where to connect the resistor of the compensation network to.

Harry Dymond has pointed out in his paper that the resistor of the compensation network can be connected to ground instead of the power supply rail and with the model amplifier used in this paper, PSRR improved by doing so.

In the paper from Michael Kiwanuka, the resistor in the model amplifier similar to mine (page 67, figure 57), is connected to the corresponding power supply rail.

This is the PSRR with the TPC resistor connected to ground:

This is the PSRR with the TPC resistor connected to the positive power supply:

Conclusion for the amplifier presented here is that PSRR is more symmetric and better overall with the resistor connected to the power supply instead of ground. My conclusion is that I had better done this investigation earlier and connected the resistor to the positive power supply rail instead of ground. With the active power supply filters added, the difference becomes negligible, but the vulnerability is still present.

The correct way to connect the resistor depends on the amplifier topology. This is why the connection differs between the amplifiers presented by Harry and Michael.

PSRR with TMC

Transitional Miller Compensation

There is only a small step from two pole compensation to transitional Miller compensation: Connecting the resistor of the compensation network to the output (NFB) node. I would need to do further research about this compensation technique and its benefits. Since TPC with the resistor conected to ground degrades PSRR, I wondered whether this is also true for TMC.

It was no big surprise to see the native PSRR being similar to TPC with the resistor conneted the wrong way.

Clipping

Bob wrote an entire chapter in his book about "Clipping control and civilized amplifier behavior", where he also explains the feedback baker clamps as one of the possible solutions. I find the idea behind feedback backer clamps very clever and since I haven't seen anybody implementing this before, I need to give this concept a chance.

During clipping, a lot of very nasty things happen to an amplifier unless counter-measures are implemented.

First, parts of the circuit are driven into saturation, which just means that they are no longer functional.

Apart from this, the negative feedback loop tries to combat the extreme deviation of the output signal from the input signal and uses all the crazy high open loop gain to do this. So the control loop gets out of control entirely trying to get a partially non-functional amplifier to attempt the impossible.

To make things worse, the compensation capacitors may also saturate and this means that the amplifier needs to clear the charge stored in the compensation capacitors first in order to become stable again. More advanced compensation schemes like MIC and TPC are more likely to have bad recovery characteristics.

Clearly, this is not going to end well. I added some illustrations depicting what might happen in such situations:

Feedback baker clamps return the clipped part of the signal back to the inverting input of the amplifier and so the feedback loop actually does not clip, but stays intact and retains control of the amplifier. The output signal waveform is clipped however and together with sensing the output stage supply voltage and dynamic adjustment of the clipping threshold, saturation of the output stage can be avoided as well. In above examples, the output stage supply voltage is +/-30V and clipping occurs with roughly 2V margin to the power supply rails. The overall result is a well-behaved amplifier that tolerates overdrive without freaking out.

Build

This is how the revision 1 assembly looks like:

Example configuration

Here is an example configuration together with the folded driver module and the four pairs BJT output stage mounted to one of the radiators of the chassis. Inter-module connections are mostly complete, only missing the negative feedback connection between front end and driver module. The power supply would be connected on the upper right hand side to the output stage. The loudspeaker would be connected to the lower right hand side of the output stage and two auxiliary floating power supplies would be connected to the front end PCB. Audio input is on the upper left hand side at the place where the BNC adapter occupies the SMA connector.

Revision History

Revision 1

Unfortunately, the first revision of the PCBs lacked some features that I found out to be necessary during evaluation of the prototype on the bench.

Revision 2

Most important, I forgot to add shunt compensation to the node where the feedback Baker clamp diverts the clipped signal to. Thus, R83 and C83 were missing in the first revision. Luckily, adding R83 and C83 does not make compensation even more complicated because the values of R82 and C82 should be copied.

I added R51 to R54, which are required for stability during clipping. The mechanism behind remains unknown to me. I just figured out they are required and need to have unusual high value.

I changed the voltage offset generation of the CCS cascodes from LEDs to resistors due to concerns about stability of the cascodes. Resistors are cheaper anyway and high impedance drive of the cascodes is better for stability in any case. The first revision using LEDs did not show any such instability though.

The reference voltage generation for the second LTP cascodes was separated from the clipping voltage reference for two reasons: First, on power on the maximum VCE of the small signal transistors could be violated and second, this improves the clipping behaviour. The important point here is that the voltage offset from the power supply rails for the cascodes voltage reference is lower than the offset (from the supply rails) for the clipping threshold.

Another small upgrade was to change the Zener diode used as floating cascode reference in the input LTP. Now a string of three red LEDs generated the reference and have lower current noise than a Zener diode. Those LEDs were saved in the CCS earlier so it was a zero cost upgrade.

Since I was struggling with Miller input compensation for quite some time, I added further compensation options to the second revision: Ordinary Miller compensation, two pole compensation and bridged T compensation.

Resistor R85 was another addition to the compensation scheme. The addition resulted from a soldering mistake during experiments and seems to help with stability. I would prefer not to install R85.

For easier experimentation with compensation I added potentiometers P1 to P4 in parallel to the resistors setting compensation.

Revision 3

Revision two addressed a lot of important issues, but I connected the TPC compensation network the wrong way (see compensation vs. PSRR).

Also, capacitor C87 was added because this is mandatory for successful implementation of TPC to this amplifier topology. (See Michael Kiwanuka's paper).

Capacitor C82 of the output shunt compensation was split in two (C821 and C822) to shunt both output nodes because I found it counterintuitive to shunt only one of the outputs like Bob does it in his schematics (what should be sufficient).

Resistor R85 was added as further improvement of the TPC network to suppress the mid-band OLG peak.

The paper from Michael Kiwanuka inspired me to add differential driven cascodes to the input LTP.

I added C113 and C114 to the reference voltage generators in order to improve PSRR significantly. Also, R113 and R114 were added as precaution to lower the risk of Q105 and Q106 oscillating.

In the third revision I removed some radiators that I found to be not necessary.

Status and Outlook

Testing the module is ongoing with different driver stages and the BJT output stage.